Method and systems for receiving plural information flows in a MIMO system

ABSTRACT

In a communication system, such as a Multiple Input Multiple Output system operating in a spatial multiplexing mode, for use, e.g., in a WLAN or HSPDA device, a plurality of information flows are received via a set of receive antennas by deriving from at least some, and possibly all, of the receive antennas, respective RF signals, and producing from the RF signals thus derived, a plurality of receive signals, each receive signal to be demodulated to recover one of the information flows transmitted. The receive signals are produced as combinations of the RF signals having applied thereto relative RF phase shift weights.

FIELD OF THE INVENTION

The invention relates to arrangements for receiving plural informationflows. A possible field of use of such arrangements are so-calledMultiple Input Multiple Output (MIMO) antenna systems.

DESCRIPTION OF THE RELATED ART

MIMO systems represent a promising solution for improving the capacity(throughput) and reliability (coverage range) of wireless communicationsystems.

In a MIMO system the transmitter is equipped with n_(T) antennas and thereceiver with n_(R) antennas operating at the same time on the samefrequency. A possible transmission mode of MIMO systems is the so-calledspatial multiplexing (SM) technique based on the transmission ofdifferent data streams across the n_(T) antennas with the goal ofincreasing the overall throughput. Recent information theory resultshave revealed that a richly scattered multi-path wireless channel iscapable of providing a huge capacity. In the presence of MIMO-SMtransmission mode, the multi-path environment can be exploited bytransmitting simultaneously on the same frequency different data streamson different transmitting antennas providing a K-fold capacity increase,where K is the minimum between the number of transmitting antennas andthe number of receiving antennas, i.e. K=min(n_(T), n_(R)) with theconstraint that n_(R)≧n_(T).

A block diagram of an exemplary MIMO system operating in a spatialmultiplexing (SM) mode is shown in FIG. 1. There, a transmitter TX isshown which transmits a plurality of information flows towards areceiver RX over a channel C.

The transmitter TX can be thought of as a serial-to-parallel converter(S/P) or, equivalently, a time de-multiplexer. Supposing that everyantenna is able to carry a data signal with a throughput equal to S, theoverall throughput of the data signal x at the input of the MIMOtransmitter TX is equal to n_(T)·S, that is n_(T) times larger than thethroughput S carried by every single antenna. The spatial multiplexingeffect across the multiple transmitting antennas introduced by theMIMO-SM transmitter leads to these data streams being mixed up in theair (i.e. in the “channel” C). If n_(R)≧n_(T) the output signal y can berecovered at the receiver RX by means of suitable signal processingalgorithms. MIMO systems also offer a significant diversity advantageand thus they can improve the coverage range with respect to singleantenna systems (SISO) by exploiting both transmit and receive antennadiversity.

The propagation channel C from the transmitter TX to the receiver RX canbe modeled, for each multi-path component, by means of a channel matrixH of complex channel coefficients with size n_(R)×n_(T). The largerspectral efficiencies (high throughputs) that can be achieved with MIMOchannels are based on the assumption that a rich scattering environmentprovides independent transmission paths from each transmit antenna toeach receive antenna. Therefore, for single-user systems, a transmissionand reception strategy that exploits this structure will achieve, withthe minimum number of transmitting and receiving antennas K=min(n_(T),n_(R)), a linear increase of the transmission rate for the samebandwidth with no additional power expenditure over a single antennasystem. This capacity increase requires a scattering environment suchthat the channel matrix between transmit and receive antenna pairs hasfull rank and independent entries and that perfect estimates of itscoefficients are available at the receiver. Performance of a MIMO systemoperating in a SM mode, in terms of throughput versus signal to noiseplus interference (SINR) ratio, will thus depend on the properties ofthe channel matrix.

The exemplary SM technique considered here is based on digital signalprocessing operations that are performed by the receiver, at base-bandlevel, and, in principle, is essentially independent of theelectromagnetic characteristics of the receiving antennas (provided theyhave omni-directional radiation patterns). In the case in question, thenumber n_(R) of receiving antennas is assumed to be larger or at mostequal to the number n_(T) of the transmitting antennas or equivalentlythe number of the transmitted spatial streams. In comparison toconventional MIMO receivers with n_(R)=n_(T), those MIMO receivershaving a number n_(R) of receiving antennas higher than the number n_(T)of multiple spatial streams provide a higher performance level, in termsof throughput versus signal to noise plus interference (SINR) ratio.This entails however a cost in terms of additional complexity due to then_(R)−n_(T) additional receivers and more complex base band (BB)algorithms.

WO-A-03/073645 describes a radio communications device comprising threeor more diverse antennas and either a plurality of transmit chains or aplurality of receive chains, and wherein there are fewer transmit orreceive chains than antennas. The radio communications device isarranged to provide multiple-input multiple-output (MIMO) communicationswith the advantage that increased data rates can be achieved in additionto cost and space reduction. The antennas employed can have directionalradiation patterns with the further advantage of providing higher levelsof signal-to-interference plus noise ratios (SINR) when employed in acellular network. The radio communications device comprises a selectorarranged to select for each receive chain or for each transmit chain anyone of the antennas for use in conjunction with that receive or transmitchain as, for example, in a switched antenna selection scheme.

WO-A-06/052058 describes a method for enhancing performance of a MIMOsystem employing a space-time coding (STC) scheme, MIMO-STC, inconjunction with transmit antenna selection scheme. The transmitterincludes N transmit antennas that are in excess of the M transmitantennas required for transmitting a signal to a space channel. Thetransmitter selects the M transmit antennas among the N transmitantennas and transmits a symbol by space-time encoding the symbol. Thereceiver includes M receive antennas for receiving a signal from thespace channel so that it detects the transmitted information symbol byusing the signal received through the receive antenna and subsequentlygenerates a transmit antenna selection information for selecting Mtransmit antennas among N transmit antennas and returns the informationto the transmitter.

TWO 2008/064696, discloses a wireless communication system wherein asub-set of RF signals received from corresponding antenna elements isselected and combined into a single RF signal. The single RF signal isprocessed and demodulated in a single processing chain, which comprisesa RF phasing network for co-phasing the selected RF signals beforecombining and a processor for controlling combining and phasing in orderto obtain a single RF signal having a radio performance indicator whichsatisfies predetermined conditions.

OBJECT AND SUMMARY OF THE INVENTION

The Applicant has observed that the need exists for arrangements for useat the receiving side of e.g. a MIMO system with a number n_(R) ofreceiving antennas larger than the number n_(T) of transmitted spatialstreams wherein only n_(T) RF receivers are required, with a consequentreduction in terms of hardware complexity.

A specific object of the invention is to provide such arrangements whichcan be used advantageously e.g. in a Wireless LAN (WLAN) or HSDPA (HighSpeed Downlink Packet Access) context while being simple and thus easyand inexpensive to produce.

The object of the invention is to provide a response to that need.

According to the present invention, that object is achieved by means ofa method having the features set forth in the claims that follow. Theinvention also relates to a corresponding system as well a WirelessLocal Area Network (W-LAN) device comprising such a system.

The claims are an integral part of the disclosure of the inventionprovided herein.

An embodiment of the invention is thus a method of receiving via a setof receive antennas a plurality of information flows, the methodincluding the steps of:

-   -   deriving from at least a subset of said set of receive antennas        respective RF signals, and    -   producing from said RF signals a plurality of receive signals,        each said receive signal to be demodulated to recover one of        said information flows,

wherein said receive signals are produced as combinations of said RFsignals having applied thereto relative phase shift weights.

In an embodiment, said respective RF signals are derived from all thereceive antennas in the set.

In an embodiment, a MIMO receiver is provided which operates on thebasis of the combination, at the RF level, of the signals received atthe output of the n_(R) antennas in order to generate n_(T) RF signalsat the input of the n_(T) RF receivers.

Embodiments of the invention provide a performance level which is higherthan that of a conventional MIMO receiver with n_(T) omni-directionalreceive antennas while the extra complexity is limited to the additionalnumber n_(R)−n_(T) of antennas and to the RF combining unit.

An embodiment of the invention is suitable to be employed in thepresence of a switched beam antenna architecture where the combinationof the signals received at the output of particular directional antennascan provide benefits in terms of array gain, diversity and interferencerejection.

An embodiment of the invention can be employed in wireless systemstransmitting multiple spatial streams as for example Wireless LAN (WLAN)compliant with the standard IEEE 802.11n, Wireless MAN (WMAN) compliantwith the standard IEEE 802.16e and the HSDPA-MIMO system proposed in3GPP Release 7.

BRIEF DESCRIPTION OF THE ANNEXED DRAWINGS

The invention will now be described, by way of example only, withreference to the enclosed figures of drawing, wherein:

FIG. 1 has been already described in the foregoing,

FIGS. 2 a to 2 c show exemplary antenna configurations,

FIG. 3 is a schematic representation of a switched beam antenna system,

FIG. 4 is a schematic representation of a RF phasing circuit,

FIGS. 5 and 6 are further schematic representations of RF phasingcircuits,

FIGS. 7 and 8 are schematic representations of switched beam antennasystems, and

FIG. 9 shows an exemplary antenna arrangement with directional antennas.

DETAILED DESCRIPTION OF EXEMPLARY EMBODIMENTS

This detailed description presents an exemplary method and a relateddevice for the implementation of a MIMO-SM receiver that, whileincluding a number n_(R) of receiving antennas larger than the numbern_(T) of transmitted spatial streams, may require only n_(T) RFreceivers with a consequent reduction in terms of hardware complexity.The exemplary architecture described herein may be based on thecombination, at the RF level, of the signals received at the output ofthe n_(R) antennas in order to generate n_(T) RF signals at the input ofthe n_(T) RF receivers.

The n_(R)−n_(T) redundant antennas at the receiver may be used tocollect different versions of the n_(T) transmitted spatial streams thatcan be combined, at the RF level, with suitable weighting factors, inorder to generate an equivalent channel matrix H with good propertiesfor the transmission of multiple spatial streams.

The minimum Euclidean distance of the received constellation may be agood parameter for determining the performance of a MIMO systemoperating in SM mode. A description of the related theory is provided byR. W. Heath and A. J. Paulraj in: “Switching Between Diversity andMultiplexing in MIMO Systems” published on IEEE Transactions onCommunications, Vol. 53, No. 6, June 2005.

In the following, an exemplary MIMO-SM system with two transmittingantennas and two receiving antennas will be considered, so thatn_(R)=n_(T)=2.

In this particular example the channel matrix H has the followingexpression

$\begin{matrix}{H = \begin{pmatrix}h_{11} & h_{12} \\h_{21} & h_{22}\end{pmatrix}} & (1)\end{matrix}$where the coefficients h_(ij) with i=1,2 and j=1,2, in the case ofomni-directional receiving antennas and a propagation scenario rich ofscattering objects, are statistically independent complex zero meanGaussian processes with an envelope having a Rayleigh probabilitydensity function and unitary variance. The minimum Euclidean distance ofthe codebook (or constellation) at the receiver (i.e. the codebookconstructed when the channel operates on each codeword) may be a goodperformance indicator of a MIMO-SM system because, assuming maximumlikelihood detection, the conditional error probability, given a channelrealization, can be determined by the distance properties of thecodebook at the receiver.

If s=[s₁, s₂]^(T) denotes a codeword comprised of two QPSK symbols s₁and s₂ respectively transmitted by the first and the second antenna of a2×2 MIMO-SM system and r=[r₁, r₂]^(T) denotes the corresponding codewordof symbols received respectively by the first and the second antenna,the following relationship applies:r=H·s+n   (2)

where n=[n₁, n₂]^(T) is the contribution of the thermal noise samples n₁and n₂ at the input of the first and the second antenna, respectively.These noise samples can be assumed to be gaussian with zero mean andvariance equal to N₀. For convenience every transmitted signal codewordis assumed here to be normalized in order to have unit energy E_(s) sothat E_(s)=∥s₁∥²+∥s₂∥²=1 and the channel H is assumed to be perfectlyknown at the receiver (via training symbols).

The following description will refer to a MIMO decoder based on amaximum likelihood (ML) algorithm and performance of the MIMO-SM systemwill be assumed to be indicated by the raw Bit Error Rate (BER) as afunction of the signal-to-noise plus interference ratio (SINR) at eachreceiving antenna. Those skilled in the art will appreciate that anyother MIMO decoder, such as e.g. a MIMO decoder based onmaximum-a-posteriori algorithm, or any other performance indicator maybe used.

The paper by R. W. Heath and A. J. Paulraj already cited in theforegoing shows that the error probability on the received codeword rconditioned to a particular channel realization H, denoted asP(error/H), is upper bounded by the following expression

$\begin{matrix}{{P\left( {{error}/H} \right)} \leq {\left( {2^{M} - 1} \right){{erfc}\left( {\frac{E_{S}}{2N_{0}}{d_{\min,r}^{2}(H)}} \right)}}} & (3)\end{matrix}$where M is the overall number of bits carried by the MIMO-SM system foreach possible transmitted codeword (e.g. with M equal to 4 for a systemwith n_(T)=2 and a QPSK modulation) and d_(min,r) ²(H) is the squaredminimum Euclidean distance of the received codebook. In the particularcase of QPSK modulation and n_(R)=n_(T)=2 transmitting and receivingantennas the squared minimum Euclidean distance of the received codebookd_(min,r) ²(H) conditioned to the channel matrix H can be computed asdetailed in the following.

If one considers two transmitted codewords s _(i) and s _(j) such that s_(i)≠s _(j). The squared Euclidean distance between two possibletransmitted codewords s _(i) and s _(j) at the receiver is given by∥H·(s _(i)−s _(j))∥²

The minimum squared Euclidean distance at the receiver can be found byminimizing this difference over all possible codewords and can beexpressed as

$\begin{matrix}{{d_{\min,r}^{2}(H)} = {{\underset{i \neq j}{\min\limits_{i,j}}{\left( {{\underset{\_}{r}}_{i} - {\underset{\_}{r}}_{j}} \right)}^{2}} = {\underset{i \neq j}{\min\limits_{i,j}}{{H \cdot \left( {{\underset{\_}{s}}_{i} - {\underset{\_}{s}}_{j}} \right)}}^{2}}}} & (4)\end{matrix}$

The impact of the minimum Euclidean distance d_(min,r) ²(H) of thecodebook (or constellation) at the receiver on the performance of aMIMO-SM system in terms of raw BER as a function of the signal tointerference plus noise ratio (SINR) measured at each receiving antennacan be evaluated on the basis of the following conditionalprobabilities:Raw BER ₁ =P{error|0.0<d _(min,r) ²(H)≦0.5}Raw BER ₂ =P{error|0.5<d _(min,r) ²(H)≦1.0}Raw BER ₃ =P{error|1.0<d _(min,r) ²(H)≦1.5}Raw BER ₄ =P{error|1.5<d _(min,r) ²(H)≦2.0}Raw BER ₅ =P{error|2.0<d _(min,r) ²(H)≦2.5}

These can be obtained by conditioning the Raw BER to different values ofthe minimum squared Euclidean distance d_(min,r) ²(H) quantized overfive different intervals derived from the corresponding probabilitydensity function. For higher values of the parameter d_(min,r) ²(H),performance in terms of raw BER exhibits a significant gain in terms ofSINR with respect to the corresponding curves obtained for smallervalues of d_(min,r) ²(H).

The minimum Euclidean distance d_(min,r) ²(H) can be calculated at thereceiver by exploiting the knowledge of channel matrix H, which isestimated by means of reference sequences. The computation of equation(4) may however require a search over a large number of transmittedcodewords, which may be prohibitive for large constellations such as 16QAM or 64 QAM. Measuring the parameter d_(min,r) ²(H) at the receivermay thus turn out to be overly complex.

It is thus possible to derive an indication about the minimum squaredEuclidean distance of the constellation received from the correspondingvalue of Raw BER measured by the baseband (BB) modules of a MIMOreceiver. Moreover, by exploiting the one-to-one relationship betweenone particular value of the raw BER and the corresponding value of theBER decoded at the output of the channel decoder or, alternatively, thecorresponding value of packet error rate (PER), it is possible to derivean indirect measure of the minimum squared Euclidean distance throughthe corresponding value of PER averaged over a certain number ofreceived packets. The lower the value of PER, the higher thecorresponding value of minimum squared Euclidean distance.

Measuring MIMO receiver performance in terms of PER involves thereception of several packets and may be slower than a correspondingmeasurement of the minimum squared Euclidean distance (which inprinciple can be performed instantaneously on every packet received).Moreover, the measure of PER can be performed with a negligiblecomplexity with respect to the measure of minimum squared Euclideandistance that, on the contrary, may impact on system complexity. Thethroughput (T) that a MIMO receiver can achieve is directly related tothe PER according to the following relationshipT=T _(peak)·(1−PER)where T_(peak) is the peak throughput achievable in the absence oferrors in the received data stream. Consequently, it may also bepossible to measure MIMO receiver performance in terms of the throughput(T) achievable in a particular propagation scenario. Moreover, MIMOwireless systems usually support adaptive modulation and codingtechniques that adaptively change the employed modulation and codingscheme. A higher signal-to-interference-plus-noise ratio (SINR) at thereceiver will translate into a higher product of the modulation orderand the channel encoding rate employed and, consequently, the maximumachievable throughput T_(peak) will be higher.

If one defines the transmission mode (TM) employed as the set ofparameters, including modulation order and channel encoding rate, whichdetermines the maximum achievable throughput T_(peak), an alternativeway for measuring the performance of a MIMO receiver may be via thetransmission mode (TM) employed in a particular propagation scenario.

For a IEEE 802.11 WLAN system, the transmission mode may correspond to aparticular transmission scheme, characterized by a particular modulationscheme (QPSK, 16 QAM, 64 QAM for example) and channel encoding rate(1/2, 3/4, 5/6 for example) that determine the maximum data rate at theoutput of PHY layer (6, 12, 18, 24, 54 Mbps for example). Newtransmission modes have been introduced for a MIMO-WLAN system compliantwith the standard IEEE 802.11n. Similarly, for a UMTS system thetransmission mode may correspond to a particular value for the transportformat (TF) that determines the maximum data rate at the output of PHYlayer (e.g. 12.2, 64, 128, 384 kbps) while for a HSPDA system thetransmission mode may correspond to a particular value of the channelquality indicator (CQI) that determines the maximum data rate at theoutput of PHY layer (e.g. 325, 631, 871, 1291, 1800 kbps).

The quality of a MIMO-SM radio link perceived by a MIMO receiver can bereasonably measured by means of a quality function Q, that depends onsome physical (PHY) and MAC layer parameters such as received signalstrength indicator (RSSI), Packet Error Rate (PER), MAC throughput (T)and employed transmission mode (TM), i.e.:Q _(s) =f(RSSI,PER,T,TM)

Usually, the higher the value of Q_(s), the higher the quality of thereceived signal at application level. Those skilled in the art willappreciate that other quality indicators as indicated in the foregoingmay be used to calculate an alternative quality function.

The function Q_(s) may thus be used as a Radio Performance Indicator(RPI) to select the beams (i.e. the RF channels) and the RF phase shiftweights to be applied. Other types of Radio Performance Indicators (RPI)may be used within the framework of the arrangement described herein. Itwill however be appreciated that, while being representative of thequality of the respective RF signal, such radio performance indicatorsas e.g. the Received Signal Strength Indicator (RSSI), Packet Error Rate(PER), Signal to Interference-plus-Noise ratio (SINR), MAC throughput(T) and employed transmission mode (TM), or any combination of theaforementioned performance indicators will be non-RF, i.e. IntermediateFrequency (IF) or BaseBand (BB) indicators.

FIGS. 2 a to 2 c show some exemplary antenna configurations including anumber of receive antennas n_(R) which will be assumed to be larger thanthe number n_(T) of transmitted spatial streams (i.e. informationflows). In the following, the RF signals received at the output of then_(R) antennas will be denoted as r_(i) where i=1,2, . . . , n_(R).

Specifically, in FIG. 2 a six antennas A₁,A₂, . . . , A₆ are arranged ona line. In FIGS. 2 b and 2 c, eight antennas A₁,A₂, . . . , A₈ areplaced equidistantly on the perimeter of a square (FIG. 2 b) and theperimeter of a circle (FIG. 2 c).

For instance, an exemplary case can be considered where the number oftransmitted spatial streams n_(T) is equal to 2 (two) and the number ofreceiving antennas n_(R) is equal to 8 (eight).

In a receiving apparatus the number of RF receivers may be equal to thenumber of receiving antennas n_(R) so that the base band (BB) processingunit has, as input, n_(R) digital signals that can be exploited forimproving the system performance in terms of coverage and throughput. Inthis case, an equivalent channel matrix H can be defined as

$H = \begin{pmatrix}{h_{11}h_{12}} \\{h_{21}h_{22}} \\{h_{31}h_{32}} \\{h_{41}h_{42}} \\{h_{51}h_{52}} \\\ldots \\{h_{n_{R}1}h_{n_{R}2}}\end{pmatrix}$so that the BB receiver may be a Maximum Likelihood (ML) receivercomputing, with the knowledge of the signals r=[r₁, r₂, . . . r_(n) _(R)]^(T) received in correspondence of the transmitted unknown symbolss=[s₁, s₂]^(T), the following metrics.d ²( r,s _(i,j))=∥( r−H·s _(i,j))∥²  (5)where s _(i,j)=[s_(i), s_(j)]^(T) is a particular codeword of thetransmitted Codebook.

This first technique for exploiting the n_(R)−n_(T) redundant antennasmay require a number of RF receivers or transceivers equal to the numbern_(R) of receiving antennas with a consequent impact on the hardwarecomplexity of the receiver at both BB and RF level.

In order to exploit the n_(R)−n_(T) redundant antennas, the receiver mayselect a set of n_(T) signals {A_(i),A_(j), . . . A_(k)} obtained at theoutput of n_(T) receiving antennas and feeding the input of the n_(T) RFreceivers with the corresponding RF signals.

For exemplary purposes, one may consider the case of n_(R)=8 andn_(T)=2. In that case, an exemplary criterion for the selection of thepair (A_(i),A_(j)) of receiving antennas may involve selecting the twoantennas (A_(i),A_(j)) with the highest values of received signalstrength indicator (RSSI) measured by the BB processing unit. Inparticular, feedback signals generated by the BB processing unit may beused to control the antenna selection unit during the measurement of theRSSI from every particular beam.

A second possible criterion may involve selecting the two antennas(A_(i), A_(j)) that provide an equivalent channel matrix H

$H = \begin{pmatrix}h_{i\; 1} & h_{i\; 2} \\h_{j\; 1} & h_{j\; 2}\end{pmatrix}$with the largest squared Euclidean distance or, alternatively, with thehighest value of quality function Q_(s) in terms of throughput (T) ortransmission mode (TM).

In the following, this technique will be referred to generally asMIMO-SM with antenna selection, independently from the particularcriterion employed for the selection of the antenna pair (A_(i), A_(j)).

An approach for exploiting the n_(R)−n_(T) redundant antennas at thereceiver may be based on the generation of n_(T) signals z=[z₁, z₂, . .. z_(n) _(T) ]^(T) by linear multiplying the vector r=[r₁, r₂, . . .r_(n) _(R) ]^(T) of n_(R) received signals for a combining matrix W withn_(T) lines and n_(R) columns

$\begin{matrix}{{W = \begin{bmatrix}w_{1,1} & w_{1,2} & w_{1,3} & \ldots & w_{1,n_{R}} \\w_{2,1} & w_{2,2} & w_{2,3} & \ldots & w_{2,n_{R}} \\\; & \; & \; & \; & \; \\w_{n_{T},1} & w_{n_{T},2} & w_{n_{T},3} & \ldots & w_{n_{T},n_{R}}\end{bmatrix}}{{so}\mspace{14mu}{that}}{\underset{\_}{z} = {W \cdot \underset{\_}{r}}}} & (6)\end{matrix}$

In the exemplary case, where the number of transmitted spatial streamsn_(T) is equal to 2 and the number of receiving antennas n_(R) is equalto 8, the matrix W has 2 lines and 8 columns.

An embodiment of a possible switched beam antenna system is shown inFIG. 3. Specifically, a number of n_(R) antennas A₁,A₂, . . . A_(n) _(R)are connected to a phasing and combining network 10, which is in turnconnected to two RF receivers 20 a and 20 b. A BB processing unit 30 isthen able to generate feedback signals 40 for controlling the network10, to perform an analysis of the signals currently received from the RFreceivers 20 a and 20 b, and to select the most suitable antennas. Thoseskilled in the art will appreciate that also a dedicated control unitmay be used in order to avoid modifications of the BB processing unit30. Such a control unit may e.g. read the measurements provided by theBB processing unit 30 and control the feedback signals 40.

Generally, the received signals r=[r₁, r₂, . . . r_(n) _(R) ]^(T) can bewritten as:r=H·s+n   (7)where n=[n₁, n₂, . . . n_(n) _(R) ]^(T) is the vector of noise andinterference samples at the input of every receiving antenna withcomponents n_(i) with i=1,2, . . . n_(R) that are supposed to bespatially white complex gaussian random variables with zero mean andvariance equal to N₀. By combining equation (7) and (6) follows thatz=W·H·s+W·n=G·s+m   (8)where G is a matrix with n_(T) lines and n_(T) columns given by theproduct of the combining matrix W and the channel matrix H and m=[m₁,m₂, . . . m_(n) _(R) ]^(T) is the vector obtained by multiplying thevector of noise and interference samples n for the combining matrix W.Given a certain channel matrix H the basic idea consists in selectingthe combining matrix W in order to obtain an equivalent channel matrix Gwith good properties in terms of minimum Euclidean distance of thereceived codebook d_(min,r) ²(G) or alternatively with higher value ofquality function Q, in terms of throughput (T) or transmission mode(TM).

Moreover, when adopting this approach, by introducing some constraintson the values of the coefficients of the combining matrix W, thecomputation of the n_(T) signals z=[z₁, z₂, . . . z_(n) _(T) ]^(T) canbe directly performed at the RF level with consequent savings in termsof hardware complexity; in this particular case, the number of requiredRF receivers is only equal to n_(T).

Assuming that every coefficient w_(i,j) with j=1,2, . . . n_(R) andi=1,2, . . . n_(T) of W has unitary module and phase equal to φ_(i,j)the product of the vector r=[r₁, r₂, . . . r_(n) _(R) ]^(T) by aparticular line w _(i)=[w_(i,1), w_(i,2) w_(i,3) . . . w_(i,n) _(R)]^(T) of the matrix W may be implemented by means the circuit shown inFIG. 4.

Specifically, such a circuit may include a set of RF phasing networks12, which are connected to the respective antennas A_(i),A₂, . . .A_(nR) and to a common combiner 14.

In the exemplary case, the phasing and combining network 10 of FIG. 3could be implemented by two of these circuits, which would then providethe signals to the RF receivers 20 a and 20 b.

A further simplification of the RF phasing network of FIG. 4 can beobtained by assuming that the phase φ_(i,j) of the weighting coefficientw_(i,j) can assume only particular quantized values.

Assuming that the phase φ_(i,j) of the coefficient w_(i,j) can takevalues in the set {0, π/2, π, 3/2 π}, the corresponding multiplicationof the received signal r_(i) for the coefficient w_(ii) can be obtainedby means of the circuit shown in FIG. 5, including a number of RF delaylines 52, 54, 56, 58 with different lengths.

It will be appreciated that, for the purposes of this description, aunitary real coefficient w_(i,j) with φ_(i,j) equal to zero will in anycase be considered as a particular case for a phase shift weight.

In a corresponding embodiment as shown in FIG. 5, the “delay” line 52will thus be a line avoiding (i.e. exempt of) any phase shift, while thedelay lines 54, 56 and 58 generate phase shifts of 90°, 180° and 270°,respectively.

An arrangement including six RF switches SW₁, SW₂, . . . SW₆ willpermit, by adequately setting the switches, to selectively obtain anyone of the four values of phase shift in the set {0, π/2, π, 3/2π}.

The exemplary processing arrangement just described thus includes atleast one RF delay line 54 to 58 to apply a phase shift weight (W) to arespective RF signal r₁, . . . , r_(nR) derived from the receiveantennas. In the embodiment shown, the processing arrangement thusincludes at least two propagation paths 52 to 58 for the RF signal r₁, .. . , r_(nR). At least one of these propagation paths 52 to 58 includesa said delay line (this is the case for the paths 54 to 58) with adifferent delay value. The switching elements SW1 to SW6 are operable toselectively direct the respective RF signal r₁, . . . , r_(nR) to thepropagation paths 52 to 58 in order implement a different phase shiftweight. One of the propagation paths, namely the path indicated byreference numeral 52 is exempt of any delay line (i.e. implements aphase shift weight equal tozero).

Implementing RF delay lines providing a specified phase shift and RFswitches for selective connection thereof is well known in the art,which makes it unnecessary to provide a more detailed descriptionherein.

The RF multiplier circuits of FIG. 4 can be simplified by assuming thatthe phase φ_(i,j) of the weighting coefficient w_(i,j) can assume onlytwo particular quantized values in the set {0, π}. The correspondingmultiplication of the received signal r_(i) for the coefficient w_(i,j)can thus be obtained by means of the circuit shown in FIG. 6.

Specifically, in this arrangement only the delay lines 52 (with no phaseshift proper) and 56 and two switches SW₁ and SW₂ are required to obtainthe RF multiplication.

A simplification of the overall receiver architecture can be obtained bysupposing that, in every line of the combining matrix W, only n_(T)−1coefficients w_(i,j) have unitary module and phase φ_(i,j) quantizede.g. over 4 or 2 different values, one coefficient is equal to 1, andthe remaining n_(R)−n_(T) coefficients are equal to zero.

In this particular case an additional constraint may be introduced byrequiring that, in every column of the combining matrix W only onecoefficient w_(i,j) with j=1,2, . . . , n_(R) has a module equal to 1.This means that each one of the n_(R) signals received contributes onlyonce to the combination.

For example, the combining matrix W may have the following structure

$W = \begin{bmatrix}{0,} & {0,} & {0,} & {w_{1,A},} & {0,} & {0,} & {w_{1,B},} & 0 \\{0,} & {w_{2,C},} & {0,} & {0,} & {0,} & {w_{2,D},} & {0,} & 0\end{bmatrix}$with the following conditions for the four coefficients different fromzero:w_(1,A)=1w_(2,C)=1w _(1,B) =w ₁=exp(jφ ₁) with φ₁ε{0, π} or

-   -   φ₁ε{0,π/2, π, 3/2π}        w _(2,D) =w ₂=exp(jφ ₂) with φ₂ε{0, π} or    -   φ₂ε{0, π/2, π, 3/2π}

Specifically, the positions of the coefficients w_(1,A) and w_(1,B) inthe first line of the combining matrix W determine, among the n_(R)=8signals r=[r₁, r₂, . . . r_(n) _(R) ]^(T) received from the antennasA₁,A₂, . . . A₈, those signals r_(A) and r_(B) that are combined throughthe RF multiplication for the weighting coefficient w_(1,B) in thefollowing denoted as w₁.

In a similar way the position of the coefficients w_(2,C) and w_(2,D) inthe second line of the combining matrix W determines the signals r_(C)and r_(D) that are combined through the RF multiplication for theweighting coefficient W_(2,D) in the following denoted as w₂.

Finally, the constraint requiring that in every column of the combiningmatrix W is only one coefficient w_(i,j) with j=1,2, . . . , n_(R) thathas a module equal to 1, corresponds to combining, at RF level, two RFsignals r_(A) and r_(B), whose weighted sum feeds the first RF receiver,that are different from the corresponding two RF signals r_(C) and r_(D)that are combined at RF level and whose weighted sum feeds the second RFreceiver.

FIG. 7 shows schematically a possible embodiment of a switched beamantenna system for the exemplary case of n_(R)=8 and n_(T)=2.Specifically, the signals r=[r₁, r₂, . . . r_(n) _(R) ]^(T) receivedfrom the antennas A₁,A₂, . . . A₈ are connected to a switching network122, which provides the signals r_(A), r_(B), r_(C) and r_(D). Theswitching network 122 may be set e.g. through the BB processing circuit30, which analyses the quality function Q_(s) and provides theinformation about the signals r_(A), r_(B), r_(C) and r_(D) which areselected.

The signals r_(A) and r_(B) are then processed by multiplying them bythe respective coefficients of the matrix W, to be then combined in acombiner 14 a and provided to the first RF processing chain 20 a.Specifically, no multiplication is necessary for the signal r_(A),because the coefficient w_(1,A) is equal to 1. Instead the signal r_(B)is multiplied with the coefficient w₁ (i.e. w_(1,B)) by a first RFphasing network 124 a.

Similarly, only the signal r_(D) may be multiplied with the coefficientw₂ (i.e. w_(2,D)) by a second RF phasing network 124 b, and the weightedsignals are combined in a combiner 14 b and provided to the second RFprocessing chain 20 b.

Applicants verified that this condition ensures that the equivalentchannel matrix G has good properties in terms of minimum Euclideandistance of the received codebook and consequently also in terms of interms of throughput (T).

The operations performed by the MIMO wireless receiver or transceiverwith redundant antennas shown in FIG. 7 are therefore the following:

-   -   determine, among the n_(R)=8 received signals, the 4 signals        r_(A), r_(B), r_(C) and r_(D) according to a first criteria, and    -   determine the values of the phases of the 2 weighting        coefficients w₁ and w₂ according to a second criteria.

The final goal is to maximize a certain quality function Q_(s) that canbe measured by the receiver in terms, for example, of received signalstrength indicator (RSSI), Packet Error Rate (PER), MAC throughput (T)and employed transmission mode (TM) or in terms of a suitablecombination of the aforementioned performance indicators so that thefirst criteria for the selection of the signals r_(A), r_(B), r_(C), andr_(D) together with the second criteria for the selection of theweighting coefficients w₁ and w₂ should be chosen with the goal ofmaximizing a quality function Q_(s).

Exemplary embodiments of criteria for the selection of the signalsr_(A), r_(B), r_(C), and r_(D) are provided in the following. In theparticular case of propagation scenarios without interference from theneighboring cells, where thermal noise is the main limiting factor, itis possible to select the four signals r_(A), r_(B), r_(C) and r_(D)with the higher values of received signal strength indicator (RSSI)measured by the BB processing unit 30.

On the contrary, for propagation scenarios with a high level ofinterference it is possible to select the 4 signals r_(A), r_(B), r_(C)and r_(D) with the highest value of signal-to-noise plus interferenceratio (SINR) measured by the BB processing unit 30. The signal-to-noiseplus interference ratio (SINR) can be measured, for example, asdifference of subsequent measures of RSSI obtained first on the usefultransmitter and then on the interfering transmitter. This approach isnot very precise when the transmissions of the reference beacon channelsof the useful and of the interfering transmitters present a certainoverlap in time.

Alternatively, it is possible to select the four signals r_(A), r_(B),r_(C) and r_(D) providing the higher values of throughput (T) measuredby the BB processing unit 30.

After having selected the four signals r_(A), r_(B), r_(C) and r_(D) itis possible to optimize the values of the coefficients w₁ and w₂ throughan exhaustive search driven by a certain performance indicator providedby the BB processing unit 30 such as the throughput (T) of the radiolink.

In case of coefficients w₁ and w₂ quantized over two different valuesfour different values of the aforementioned performance indicators arecomputed, while, in the case of coefficients quantized over fourdifferent values, sixteen different values of the performance indicatorare computed and the coefficients w₁ and w₂ providing the highest valueof the performance indicator are selected.

Another exemplary approach involves selecting the four signals r_(A),r_(B), r_(C) and r_(D) jointly with the four or sixteen 16 values of thecoefficients w₁ and w₂ through an exhaustive search driven by a certainperformance indicator provided by the BB processing unit 30, such as thethroughput (T) of the radio link. This approach may require a longercomputational time but provides the optimal combination of the signalsreceived and the weights that maximize the performance indicatorconsidered.

In the particular case where the number of transmitted spatial streamsn_(T) is equal to two and the number of receiving antennas n_(R) isequal to eight, the overall number of combination of the signals r_(A),r_(B), r_(C) and r_(D) with the values of the coefficients w₁ and w₂ isequal to 6720 for coefficients w₁ and w₂ quantized over 2 values and to26880 for coefficients w₁ and w₂ quantized over 4 values. Under theseassumptions an exhaustive search may not be feasible for mostapplications.

Possible simplifications of the switching network 122 for selecting thesignals r_(A), r_(B), r_(C) and r_(D) can be envisaged in order toreduce the time for performing the exhaustive search.

In FIG. 8 shows an exemplary simplified switching network where onlysome particular combinations of the signals received by the differentbeams can be provided to the two RF processing chains 20 a and 20 b.

Specifically, the switching network 122 of FIG. 7 may be implementede.g. by means of four switches. A first switch 122 a may select thesignal r_(A) among the signals provided from the antennas B₆ and B₈.Similarly the switches 122 b, 122 c, 122 d may select the signal r_(B),r_(C) and r_(D) among the signals provided from the antennas B₂ and B₄,B₅ and B₇, and B₁ and B₃, respectively.

In the exemplary implementation of the switching network shown in FIG.8, the overall number of combination of the signals r_(A), r_(B), r_(C)and r_(D) with the values of the coefficients w₁ and w₂ is equal to 64for coefficients w₁ and w₂ quantized over 2 values and to 256 forcoefficients w₁ and w₂ quantized over 4 values so that the searchprocedure is greatly simplified at the cost of a slight reduction of thesystem performance due to non exhaustive search procedure of thereceived signals.

The technique proposed for the reception of multiple spatial streamswith redundant antennas can be employed in the presence of receivingantennas with omni-directional radiation patterns or alternatively inthe presence of directional antennas with the further advantage ofintroducing a beneficial effect of spatial filtering (through theselection of four out of eight directional receiving antennas) that canincrease system performance in the presence of a propagation scenariolimited by interferers that are not spatially white.

The most common antenna types for WLANs have omni-directional radiationpatterns. Omni-directional antennas propagate RF signals in alldirections equally on a horizontal plane (azimuth plane). The gainachieved with an omni-directional antenna can somehow not be sufficientto reach certain coverage ranges. Higher gain values can be obtainedwhen adopting a directional antenna, which is able to focus thetransmitted and received RF energy in a particular direction thusachieving higher coverage ranges.

Moreover, in order to achieve an improvement both in terms of coverageand throughput, the spatial domain of the propagation environment can beexploited by adopting multiple antennas. Such systems increase theinformation available at the receiver end by means of appropriate signalprocessing techniques, thus reducing the impairments such as multi-pathinterference introduced during the transmission over the propagationchannel. The exploitation of multiple directional antennas can lead togood performance in terms of throughput and coverage range.

In the particular case of directional antennas, the design of theantenna system has to take into account that the received signals arrivefrom all possible directions. In particular the Angle of Arrival (AoA)in the azimuth plane may take all the possible values between 0 and 360degrees due to the presence of many scattering objects surrounding thereceiver.

The AoA distribution in the elevation plane depends on the transmitterposition. The transmitters are generally placed in the centre of theroom or fixed to a wall or to the ceiling in order to provide maximumcoverage. It is then reasonable to assume the AoA in the elevation planeis concentrated around the horizontal direction with an angular spreadlower than 180 degrees.

The top view of a possible multiple directional antenna system is shownin FIG. 9 in the particular case of N=8 directional antennas B₁, B₂, . .. B₈ (placed on the vertexes of a regular octagon circumscribed by acircumference).

Experimental results have been obtained by the Applicant, i.a. withreference to three different receiver architectures, namely:

-   -   MIMO 2×2: the basic reference system, equipped with 2        omni-directional antennas that feed 2 RF receivers at the        receiver end,    -   MIMO 2×8 with selection of 2 directional antennas, where the        receiver selects (among the 8 available receiving antennas) a        suitable pair of antennas and feeds their signals to the input        of 2 RF receivers. The selection of the 2 antennas is carried        out by referring to the two highest Received Signal Strength        Indicator (RSSI) values, and    -   MIMO 2×8 with RF combination of 4 directional antennas, where        the receiver selects (among the 8 available receiving antennas)        two pairs of signals feeding the input of the 2 RF receivers.        The 4 signals are selected as follows. First the selection of 4        antennas is carried on by determining, among the 8 signals        received by the available antennas, the 4 signals that maximize        a certain performance indicator (e.g. RSSI in a noise-limited        scenario, SINR in an interference-limited scenario).        Subsequently the receiver determines the appropriate phase        values of the 2 weighting coefficients that maximize the same        performance indicator value for the combined signals. The        architecture is shown in FIG. 8.

Four different propagation scenarios were used to evaluate systemperformance. These four scenarios refer to different propagationenvironments that are all characterized by the presence of a transmitter(TX) position and a receiver (RX) position, by two separate clusters ofscatterers. All the scenarios had the same Angle of Arrival (AoA) andAngle of Departure (AoD) values but have different Angle Spread (AS)values. The angle spread values were subsequently reduced from Case 1 toCase 4, this leading to less scattered environments thus achievinghigher correlation conditions. In the two first scenarios a richlyscattered propagation leads to low correlation, and in the last casepoorly scattered propagation leads to high correlation.

Results were obtained, in the particular case of n_(R)=8 and n_(T)=2, byemploying the directional antenna system as shown in FIG. 9 and thearchitecture of the MIMO wireless transceiver with redundant antennasshown in FIG. 8.

Performance of these two 2×8 MIMO antenna systems were compared withthat of a conventional 2×2 MIMO antenna system, with specific referenceto a propagation scenario where the limiting factor is the interferencegenerated by the other users (access point or clients), which wereassumed to be uniformly distributed in the spatial domain (spatiallywhite).

The results demonstrate that the enhancement in performance, expressedin terms of raw BER, is approximately 6 dB in the range of interest.This value is slightly reduced (to 5 dB) when the propagation conditionsare such that the reduced angle spread leads to higher correlation ofthe received signals.

The results show an even higher gain in performance in the case ofspatially colored interferers.

Without prejudice to the underlying principles of the invention, thedetails and the embodiments may vary, even appreciably, with referenceto what has been described by way of example only, without departingfrom the scope of the invention as defined by the annexed claims.

1. A method of receiving via a set of receive antennas a plurality ofinformation flows, wherein the number of said received antennas ishigher than the number of said information flows, comprising: derivingfrom at least a subset of said set of receive antennas respective RFsignals; selecting among said RF signals different sets of selected RFsignals; producing a plurality of receive signals as combinations of theselected RF signals in said different sets, having applied theretorelative RF phase shift weights, each said receive signal to bedemodulated to recover one of said information flows in said plurality,wherein said different sets of said selected RF signals each include arespective set of RF signals, whereby each said RF signal is included inonly one of said different sets of RF signals.
 2. The method of claim 1,comprising deriving respective RF signals from all of the receiveantennas in said set.
 3. The method of claim 1, comprising: generatingfor said RF signals at least one non-RF radio performance indicatorrespective of the quality of said RF signals; and generating said RFphase shift weights as a function of said at least one non-RF radioperformance indicator.
 4. The method of claim 3, wherein said at leastone radio performance indicator is selected from: received signalstrength indicator, packet error rate, signal to interference-plus-noiseratio, MAC throughput and employed transmission mode, and combinationsthereof.
 5. A system for receiving via a set of receive antennas, aplurality of information flows, wherein the number of said receiveantennas is higher than the number of said information flows,comprising: a connection arrangement configured to derive from at leasta subset of said set of receive antennas respective RF signals, and forselecting among said RF signals different sets of selected RF signals;and a processing arrangement including a RF combining network configuredto produce a plurality of receive signals as combinations of theselected RF signals in said different sets, having applied theretorelative RF phase shift weights, each said receive signal to bedemodulated to recover one of said information flows in said plurality,wherein said connection arrangement is configured for selecting saiddifferent sets of said selected RF signals to include each a respectiveset of RF signals, whereby each said RF signal is included in only oneof said different sets of RF signals.
 6. The system of claim 5,comprising said connection arrangement being configured to deriverespective RF signals from all of the receive antennas in said set. 7.The system of claim 5, comprising a configuration configured to:generate for said RF signals at least one non-RF radio performanceindicator representative of the quality of said RF signals; and generatesaid RF phase shift weights as a function of said at least one non-RFradio performance indicator.
 8. The system of claim 7, wherein said atleast one radio performance indicator is selected from: received signalstrength indicator, packet error rate, signal to interference-plus-noiseratio, MAC throughput and employed transmission mode, and combinationsthereof.
 9. The system of claim 5, wherein said connection arrangementcomprises inputs equal in number to the number of said receive antennasand said processing arrangement comprises outputs equal in number tosaid information flows.
 10. The system of claim 5, wherein saidprocessing arrangement comprises at least one RF delay line configuredto apply a respective RF phase shift weight to one of said selected RFsignals in said different sets.
 11. The system of claim 10, wherein saidprocessing arrangement comprises: at least two propagation paths forsaid respective one of said selected RF signals in said different sets,wherein at least one of said propagation paths comprises one said delayline; and associated switching elements configured to selectively directsaid respective one of said selected RF signals to said at least twopropagation paths.
 12. The system of claim 11, wherein one of said atleast two propagation paths is exempt from delay lines.
 13. A wirelesslocal area network device comprising a wireless communication systemaccording to claim
 5. 14. A high speed downlink packet access devicecomprising a wireless communication system according to claim 5.